In many applications a power converter is required to provide a voltage within a predetermined range formed from a voltage source having a different voltage level. Some circuits are subject to uncertain and undesirable functioning and even irreparable damage if supplied power falls outside a certain range. More specifically, in some applications, a precise amount of power is required at known times. This is referred to as regulated power supply.
In order to control a power converter to deliver a precise amount of power as conditions require, some form of control of the power converter is required. This control can occur on the primary side of an isolation transformer or the secondary side. A closed loop feedback control system is a system that monitors some element in the circuit, such as the circuit output voltage, and its tendency to change, and regulates that element at a substantially constant value. Control on the secondary side of a power converter can use a monitored output voltage as feedback control, but requires the use of some communication from the secondary to the primary side of the isolation transformer to control the primary side switching element. Control on the primary side can readily control the primary side switching element, but requires some feedback mechanism from the secondary side to the primary side to convey the status of the monitored element. In some applications, an optical coupler circuit, or opto coupler, is used to transmit feedback signals while maintaining electrical isolation between the primary and secondary sides.
FIG. 1 illustrates a conventional regulated switch mode power converter including an optical coupler circuit. The power converter 2 is configured as a traditional flyback type converter. The power converter 2 includes an isolation transformer 4 having a primary winding P1 and a secondary winding S1. The primary winding P1 is electrically coupled to an input voltage Vin and a driving circuit including a transistor 8, a resistor 12, and a controller 10. A capacitor 28 is coupled across the input Vin and coupled with the primary winding P1. Input voltage to the circuit may be unregulated DC voltage derived from an AC supply after rectification and filtering. The transistor 8 is a fast-switching device, such as a MOSFET, the switching of which is controlled by the fast dynamic controller 10 to maintain a desired output voltage Vout. The controller 10 is coupled to the gate of the transistor 8 and provides a pulse width modulated (PWM) switching signal as the driving signal. As is well known, DC/DC conversion from the primary winding P1 to the secondary winding S1 is determined by the duty cycle of the PWM switching signal provided to the transistor 8. The secondary winding voltage is rectified and filtered using the diode 6 and the capacitor 22. A sensing circuit and a load 14 are coupled in parallel to the secondary winding S1 via the diode 6. The sensing circuit includes resistor 16, resistor 18, and a secondary controller 20. A secondary controller 20 senses the output voltage Vout across the load.
In this configuration, the power converter is controlled by driving circuitry on the primary side, and the load coupled to the output is isolated from the control. As such, a monitored output voltage used for voltage regulation is required as feedback from the secondary side to the control on the primary side. The power converter 2 has a voltage regulating circuit that includes the secondary controller 20 and an optical coupler circuit. The optical coupler circuit includes two galvanically isolated components, an optical diode 24 coupled to the secondary controller 20 and an optical transistor 26 coupled to the controller 10. The optical diode 24 provides optical communication with the optical transistor 26 across the isolation barrier formed by the transformer 4. The optical coupler circuit in cooperation with the secondary controller 20 provides feedback to the controller 10. The controller 10 accordingly adjusts the duty cycle of the transistor 8 to compensate for any variances in an output voltage Vout. The feedback provided to the controller 10 typically represents an error determined by the controller 20. A comparator within the controller 20 compares the output voltage Vout sensed when the transistor 8 is OFF to a reference voltage, the difference, or error, is provided as the feedback. The error is used to adjust the pulse width, or duty cycle, of the PWM switching signal provided to the transistor 8. If the sensed output voltage Vout is below the reference voltage, then the pulse width is increased to provide additional power to the output. If the sensed output voltage Vout is above the reference voltage, then the pulse width is decreased to reduce power to the output. The sign of the error determines whether the pulse width is increased or decrease, and the magnitude of the error determines by how much the pulse width is increased or decreased.
However, the use of an optical coupler circuit in and of itself presents issues. Firstly, the optical coupler circuit adds extra cost. In some applications, the optical coupler circuit can add more cost to the power converter than the isolation transformer. The optical coupler circuit also adds to manufacturing and testing costs. Furthermore, the performance of the optical coupler circuit degrades over time and therefore introduces another potential point of failure in the overall power converter. Also, characteristics of the optical coupler circuit must be accounted for in the overall circuit design. For example, the optical diode component is non-linear and as such a correlation between the optical diode and the optical transistor must be established. The optical coupler circuit also has delays related to the operation of the optical diode and the optical transistor, and the operation of the optical diode requires a well defined DC level. As a result, it is generally desirable to avoid the use of an optical coupler circuit.
A next generation of feedback control does not use optical control circuitry. Instead, the transformer is used to convey real-time feedback signaling from the secondary side to the primary side. In such an application, the transformer includes an auxiliary winding on the primary side that is magnetically coupled to the secondary winding. FIG. 2 illustrates a conventional regulated power converter including a magnetically coupled feedback circuit. The power converter 32 is configured as a traditional flyback type converter. The power converter 32 includes an isolation transformer 34 having a primary winding P1 and a secondary winding S1. The primary winding P1 is electrically coupled to an input voltage Vin and a driving circuit including a transistor 44, a resistor 46, and a controller 42. A capacitor 58 is coupled across the input Vin and coupled with the primary winding P1. Input voltage to the circuit may be unregulated DC voltage derived from an AC supply after rectification and filtering. Similar to the power converter in FIG. 1, the transistor 44 is a fast-switching device controlled by the fast dynamic controller 42 to maintain a desired output voltage Vout. The secondary winding voltage is rectified and filtered using the diode 36 and the capacitor 38, with the output voltage Vout delivered to the load 40.
The power converter 32 has a feedback loop that includes a magnetically coupled feedback circuit coupled to the secondary winding S1 of the transformer 34 and the controller 42. The magnetically coupled feedback circuit includes a diode 48, a capacitor 50, resistors 52 and 54 and an auxiliary winding 56. The auxiliary winding 56 is coupled in parallel to the series of resistors 52 and 54.
The auxiliary winding 56 is also magnetically coupled to the secondary winding S1. When the current through the diode 36 is zero, the voltage across the secondary winding S1 is equal to or proportional to the voltage across the auxiliary winding 56 depending on the turns ratio. The voltage divider formed by the resistors 52 and 54 can be configured to match the turns ratio of the secondary winding S1 and the auxiliary winding 56 so that the voltage across the secondary winding S1 equals the voltage VA. This relationship provides means for communicating the output voltage Vout as feedback to the primary side of the circuit. The voltage across the auxiliary winding 56 is measured when it is determined that the current through the diode 36 is zero, which provides a measure of the voltage across the secondary winding S1 and therefore the output voltage Vout.
The voltage VA is provided as a feedback voltage VFB to the controller 42. The current through the transistor 44 is also provided as feedback current IFB to the controller 42. The controller 42 includes a real-time waveform analyzer that analyzes input feedback signals, such as the feedback voltage VFB and the feedback current IFB. Similarly to the controller 10 in FIG. 1, the controller 42 uses the feedback signals to adjust a duty cycle of the PWM signal that drives the transistor 44.
In general, control intricacies of the controller 42 are aligned with control argument sampling to achieve overall system functional performance. Sampling argument is in the form of current, voltage and impedance. System functional performance is in the form of pulse width modulation (PWM), pulse frequency modulation (PFM) and pulse amplitude modulation (PAM). The feedback signal received by the controller 42 requires some status integrity, such as no noise on the DC level, no disturbance on the switching waveform and to some degree represent a combination of analog and digital representations. The voltage across the auxiliary winding 56 typically forms a pulse train with frequency corresponding to the switching frequency of the main transistor 44. The voltage across the auxiliary winding 56 when the secondary winding current is zero, which corresponds to the diode 36 current equaling zero, corresponds to the falling edge of the pulse. As such, measuring an accurate voltage value requires that the pulse is well defined with sufficient pulse integrity particularly at the falling edge. Further, the voltage value immediately following the rising edge includes ringing due to the leakage impedance of the transformer. As such, pulse integrity also requires sufficient time for the voltage value to stabilize following the rising edge. Higher switching frequencies minimize the pulse width and therefore provide less time for voltage stabilization. For at least these reasons, providing a pulse with sufficient pulse integrity is often difficult to achieve.
The topologies of the conventional regulated switch mode power converters shown in FIGS. 1 and 2 limit the efficiency due in part to switching losses, conduction losses, and leakage impedance losses. In an effort to reduce or eliminate the switching losses and reduce EMI noise the use of “resonant” or “soft” switching techniques has been increasingly employed in the art. The application of resonant switching techniques to conventional power converter topologies offers many advantages for high density, and high frequency, to reduce or eliminate switching stress and reduce EMI. Resonant switching techniques generally include an inductor-capacitor (LC) subcircuit in series with a semiconductor switch which, when turned ON, creates a resonating subcircuit within the converter. Further, timing the ON/OFF control cycles of the resonant switch to correspond with particular voltage and current conditions across respective converter components during the switching cycle allows for switching under zero voltage and/or zero current conditions. Zero voltage switching (ZVS) and/or zero current switching (ZCS) inherently reduces or eliminates many frequency related switching losses.
The application of such resonant switching techniques to conventional power converter topologies offers many advantages for high density, high frequency converters, such as quasi sinusoidal current waveforms, reduced or eliminated switching stresses on the electrical components of the converter, reduced frequency dependent losses, and/or reduced EMI. However, energy losses incurred during control of zero voltage switching and/or zero current switching, and losses incurred during driving, and controlling the resonance means, are still problematic.
Several power converter topologies have been developed utilizing resonant switching techniques, for example the co-owned U.S. Pat. No. 7,764,515 entitled “Two Terminals Quasi Resonant Tank Circuit,” to Jansen et al. (Jansen), which is hereby incorporated in its entirety by reference. Jansen is directed to a flyback type converter including a quasi-resonant tank circuit. FIG. 3 illustrates the flyback type converter of Jansen. The quasi-resonant flyback converter 60 includes a conventional flyback converter including a transformer 62, a transistor 72, a controller 70, a diode 64, a capacitor 66, and a load 68 with the addition of a quasi-resonant tank circuit formed by a transistor 76, diodes 78, 80 and 82, and capacitors 84 and 86. When the transistor 72 is turned ON, the transistor 76 is turned OFF, and the primary winding of the transformer 62 is connected to the input supply voltage such that the input supply voltage appears across the primary winding, resulting in an increase of magnetic flux in the transformer 62 and the primary winding current rises linearly. No current flows through the secondary winding of the transformer 62 because the diode 64 is reverse biased. When the transistor 72 is turned OFF, the transistor 76 turns ON parametrically, without control of a separate control circuit. The diodes 78, 80 and 82 and the capacitor 86 function as driving circuitry for the transistor 76. With the transistor 76 turned ON, the capacitor 84 is essentially coupled in parallel to the transformer 62, and the energy previously stored in the primary winding causes current to circulate in the circuit formed by the capacitor 84 and the primary winding, forming a resonant tank. As with a conventional flyback converter, energy stored in the primary winding is delivered to the load while the transistor 72 is turned OFF. However, in the quasi-resonant flyback converter 60 of FIG. 3, a portion of the resonant energy generated in the resonant tank is also delivered to the load while the transistor 72 is turned OFF and the transistor 76 is turned ON. In this manner, the quasi-resonant flyback converter 60 of FIG. 3 delivers peak energy equal to energy from the typical flyback operation plus the resonant energy. However, current flow within the resonant tank cycles between positive and negative current flow through the primary winding. The configuration of the secondary side circuit, in particular the diode 64, only allows delivery of resonant energy during one direction of primary winding current flow. Resonant energy corresponding to the other direction of primary current flow is not delivered.
In addition to providing an increase in peak energy, the quasi-resonant flyback converter of FIG. 3 provides the conventional advantages associated with a resonant circuit, such as reduced frequency dependent losses and EMI.